Two-channel fast-sequencing high-dynamics GPS navigation receiver

ABSTRACT

A GPS navigation receiver channel tracks satellites&#39; Doppler frequencies sequentially during the twenty milliseconds that the signal is coherent in each bit time of the navigation data modulation. It measures frequency by differencing the angles of signal-vector sums weighted by parabolic humps, exploiting the commutativity of linear processing operations to raise the signal-to-noise ratio before the nonlinear operation phase of detection.

This application is a divisional of application Ser. No. 08/990,495filed on Dec. 15, 1997, now U.S. Pat. No. 5,949,374.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates generally to global positioning system devices andnavigation receivers and more specifically to methods and apparatus forproviding accurate position, velocity, and time solutions inhigh-acceleration, low-signal applications using only two receiverchannels.

2. Description of the Prior Art

A unique direct-sequence spectrum-spreading code modulates each globalpositioning system (GPS) satellite signal by alternating the signal'sphase by one hundred eighty degrees. The receiver commonly despreads thesignal by multiplying it by a replica of the transmitted code. Thedespread signal is the sum of a component in phase with a real orhypothetical local-oscillator signal and of a second component ninetydegrees out of phase with that local oscillator, which componentstogether constitutes a two-dimensional signal vector whose anglecorresponds to the despread signal's phase.

A data message carrying data that describe the satellites' positions andcarry other information about the GPS system also modulates the signalby modulo-two addition with the spreading code at a fifty-hertz rate.Since the data message's content is not generally known in advance atthe receiver, the receiver design commonly assumes the transmittedsignal to be coherent for only the twenty milliseconds of each bit time.

In effect, the satellite's and the receiving antenna's relative motionfurther modulates the phase of the GPS signal, but in a continuous wayrather than in steps of one hundred eighty degrees. The antenna'srelative velocity therefore has the effect of a frequency modulation, orDoppler shift. The receiver commonly makes measurements of thismotion-caused phase modulation, either as it affects the phase of thespreading-code modulation or the phase of the despread carrier or both.

The receiver commonly tracks the phase of the spreading code by inducingsmall code misalignments and measuring their effect in order to maximizethe amplitude of the despread signal. The receiver also commonlymeasures or tracks the phase of the despread signal, whether or notthose phase measurements also contribute directly to the navigationsolution, in order to compute the Doppler shift, by manipulating thefrequency or phase or both of the local-oscillator signal. The Dopplershift, which is the rate of change of the signal phase, must be known totune the receiver, since the possible Doppler shift of many kilohertz ismuch greater than the narrow bandwidth needed to receive each despreadGPS signal.

The receiver commonly averages over time to reduce the noise of phasemeasurements, either before or after the nonlinear phase-detectionoperation of measuring the signal vector's phase angle. Because of thisoperation's nonlinearity, the signal must dominate the noise at thephase detector if the receiver is to recover useful phase informationfrom the signal. This effect, in which signal-to-noise ratios below acertain detection threshold at the antenna cause substantial loss ofinformation, is common to angle-modulation systems, including commonfrequency-modulation broadcasting.

Detecting phase first in a wide bandwidth to create a phase functionφ(t), then averaging over a time T which coincides with or is within thetwenty-millisecond data-bit time during which the signal is coherent$\varphi_{u} = {\frac{1}{T}{\int_{{- T}/2}^{T/2}{{\varphi (t)}\quad {t}}}}$

has the advantage of eliminating the need to synchronize anypre-detection averaging or band-limiting with the bit edges, since theduration of the contamination of the phase function by the incoherenceat each data-bit edge is a negligible fraction of the bit time. Thistechnique is therefore often used when a high signal-to-noise ratio canbe relied upon.

Averaging the signal vector first for as long a time as it is coherent,that is, for the twenty-millisecond bit time,$s_{u} = {\frac{1}{0.02}{\int_{- {.01}}^{.01}{{s(t)}\quad {t}}}}$

has the advantage of significantly lowering the detection threshold, forexample, by a substantial thirteen decibels when the predetectionaveraging is lengthened from one millisecond to the full twentymilliseconds. This allows use of much noisier signals. Although thescaling shown is correct for the vector average, the scaling of this orany predetection averaging is actually irrelevant since phase detectiondiscards amplitude information.

Except in special cases in which time, altitude, or other positioninformation is known, a GPS navigation receiver must track at least foursatellites to find its antenna's position in three dimensions. Areceiver that has enough channels can dedicate a channel to trackingeach satellite. When there are more satellites to track than channels totrack them, receivers ordinarily resort to time-sharing strategies.Deliberately providing only a few channels to be shared among thesatellites tracked reduces size, hardware complexity, and cost, at theexpense of loss of signal power or risk of loss of carrier-phase lock.

A “multiplexing” receiver, as referred to herein, completes at least onecycle of channel-sharing during each twenty-millisecond bit time. Itoperates otherwise much like a multi-channel receiver, receiving phaseand data almost continuously from all satellites tracked. However, afour-satellite multiplexing receiver averages the signal for no morethan one-fourth of the bit time, and therefore pays a six-decibeldetection-threshold penalty.

A “sequencing” receiver, as referred to herein, dwells on each satellitefor one or more bit times. Sequencing receivers usually include twochannels, one to sequence among the satellites for navigation and one todwell on one satellite for a longer time to accumulate data, since thesequencing channel misses most of the data bits from any one satellite.

A sequencing receiver that dwells on each satellite for the one bit timeneeded for optimum linear processing is a “fast-sequencing” type, asused herein. Assuming that it can visit four satellites within onehundred milliseconds, the largest step that a nineteenmeter-per-second-per-second acceleration can produce in that100-millisecond cycle time is ninety-five millimeters or one-halfwavelength at the L1 carrier frequency of 1.5754 gigahertz. Thusacceleration of 1.94 times that of earth's gravity, or 1.94 G, is neededbefore the phase uncertainty exceeds the maximum that can be identifiedunambiguously.

This does not mean however that it is possible to track carrier phaseunder two-G acceleration with a single GPS receiver channel. Even thoughthe carrier phase can be determined in a single 100-millisecondinterval, the Doppler shift due to velocity is still unknown. So,without a way to remove the effect of the present velocity from the nextmeasurement, the phase-shift due to further acceleration will add tothat due to present velocity. The quickest way to measure velocity fromphase measurements every one hundred milliseconds is by differencingsuccessive phase measurements to get the average velocity between them.The greatest velocity change that a 0.97 G or 9.5meter-per-second-per-second acceleration can produce in the one hundredmilliseconds between the centers between velocity measurements is 950millimeters per second, which velocity corresponds to a phase change ofninety-five millimeters in the one hundred milliseconds between phasemeasurements or one-half wavelength at the L1 frequency. The practicalresult is to preclude tracking carrier phase per se across the gaps ineach satellite's reception except for the most sedate applications withaccelerations under one G.

Consequently, sequencing receivers have commonly been designed toreacquire each satellite for each new cycle of the sequence. Such areceiver ordinarily needs about five seconds to sequence a channelaround to all the satellites being tracked. Thus it is a“slow-sequencing” type as used herein. The signal strength needed toacquire or reacquire a satellite is much greater than that needed tomaintain continuous tracking. Thus the signals must be strong; soconventional slow-sequencing receivers, like multiplexing receivers, areparticularly hampered by low signal levels. In summary, slow-sequencingreceivers sequence slowly because they must reacquire the satellites;and they must reacquire the satellites because so much time elapsesbetween successive measurements due to their slow sequencing.

SUMMARY OF THE PRESENT INVENTION

It is therefore an object of the present invention to provide a GPSnavigation receiver that uses fewer receiver channels.

It is another object of the present invention to provide a GPSnavigation receiver that is useful in high acceleration environments.

It is another object of the present invention to provide a GPSnavigation receiver that is useful in low signal-to-noise environments.

Briefly, a GPS navigation receiver embodiment of the present inventioncomprises one receiver channel for position fixing and another receiverchannel for acquisition and for reading the satellite data. The formertracks each satellite for the entire duration of the twenty millisecondsthat the signal is coherent in one bit time of the navigation datamodulation, but it tracks in terms of frequency rather than phase.

An advantage of the present invention is that a navigation satellitereceiver is provided that can sequence through all the necessary foursatellites in five bit times or 100 milliseconds without necessitatingany satellite reacquisitions and thereby operate reliably in otherwiseunfavorable signal environments.

Another advantage of the present invention is that a high-dynamicsnavigation satellite receiver is provided that needs only two channels,one for tracking and the other for acquisition, for accelerations up toabout ten times that of gravity.

These and other objects and advantages of the present invention will nodoubt become obvious to those of ordinary skill in the art after havingread the following detailed description of the preferred embodiment thatis illustrated in the drawing figures.

IN THE DRAWINGS

FIG. 1 is block diagram of a fast-sequencing navigation satellitereceiver embodiment of the present invention; and

FIG. 2 is a graph that represents the sequencing method used by thereceiver of FIG. 1.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

FIG. 1 illustrates a fast-sequencing navigation satellite receiverembodiment of the present invention, referred to herein by the generalreference numeral 10. The receiver 10 comprises an antenna 12 forreceiving L-band radio transmissions from orbiting navigationsatellites. Assuming that there is no independent position or timeinformation available to the receiver 10, a minimum of four suchsatellites must be visible to the antenna 12 in order to provide timeand three dimensional position and velocity solutions. A radio frequency(RF) amplifier 14 boosts the radio signals for down-conversion anddemodulation. A first receiver channel 16 sequences among the visiblesatellites to collect twenty consecutive milliseconds worth ofmeasurements from each before moving to sample the next. A secondreceiver channel 18 provides for the reception of the navigation datamodulation and for searching for new satellites to track.

A pair of correlators 20 and 22 despreads the signal, allowingnarrow-band filtering to provide a processing gain that lifts thereceived signal out of the noise. A pair of samplers 24 and 26 digitizethe signals for processing by a digital signal processor (DSP) 28. Thefirst receiver channel 16, while tracking the satellites' despreadcarriers sequentially, also tracks the spreading code phase in aconventional manner. The second receiver channel 18 tracks one satellitecontinuously in any conventional manner. A navigation computer 30computes the receiver's position, velocity, and time from the apparentdistances, or pseudo-ranges, between the satellites and the receiver,based on the known times of code transmission and the presumed time atthe receiver, measured through the first receiver channel 16 and fromthe satellite ephemeris information decoded through the second receiverchannel 18 by the DSP 28.

The first receiver channel 16 sequences among several satellites anddwells long enough on each to track the Doppler shift by measuring theamount by which it has changed from the value known from the previousmeasurement. Such measurement allows tracking lock to be maintainedacross four or more navigation satellites.

FIG. 2 represents the method used by the first receiver channel 16 tosequence among the navigation satellites, for example, four spacevehicles (SV1-SV4). Each navigation satellite transmits fifty-baudnavigation data whose bit edges, depending on the distance betweensatellite and receiver, differ among the several satellites'transmissions, represented here in FIG. 2 as double-valued squarewavewaveforms with possible transitions every twenty milliseconds. The firstreceiver channel 16 is provided with the appropriate Doppler correctionand the correlator 20 is provided with the appropriate code and codephase to sequentially track SV1, SV3, SV2 and then SV4, for example, asrepresented in FIG. 2 by a trace SEQ. The first navigation satellite SV1is tracked continuously from the beginning of the data bit for twentymilliseconds to the end of the bit time. The navigation computer 30 thenprovides the first receiver channel 16 with the appropriate Dopplercorrection and the correlator 20 with the appropriate code and codephase to track navigation satellite SV3, which is the satellite whosenext data transition occurs soonest. A small idle period exists untilthe start of the beginning of the data bit from navigation satelliteSV3. Then measurements are taken for the twenty milliseconds to the endof the bit time. This continues for each satellite being tracked.

Because the sequenced receiver channel 18 switches to track the signalduring a whole bit time from the next satellite which has the earliestbit transition, the aggregate time wasted waiting for each completetwenty-millisecond navigation-channel data-bit time to begin on the nextsatellite can never exceed twenty milliseconds. Thus four satellites canall be visited within 100 milliseconds.

In the present invention, the sequencing receiver channel 18 measuresthe Doppler shift directly during each twenty-millisecond period thatthe signal from each satellite is coherent. The effect of presentvelocity is thus fully compensated for. Only future acceleration remainsa problem. At a minimum, each velocity measurement must provide enoughinformation to make the next velocity measurement 100 millisecondslater. Thus even if the acceleration were as much as ten G's, thevelocity uncertainty after 100 milliseconds would be 9.8 meters persecond, or 51.5 hertz Doppler shift at the L1 carrier frequency. This isnot an unreasonable frequency shift to measure in twenty milliseconds bythe method of the present invention.

The method of the present invention for measuring the Doppler shift isequivalent to measuring the rate of change of a scalar phase by fittinga straight line to the data using a least-squares criterion. This inturn is equivalent to averaging the phase measurement as a function oftime across the twenty-millisecond bit time with a ramp-like weighting:$\omega_{r} = {{\frac{12}{T^{3}}{\int_{{- T}/2}^{T/2}{{{t\varphi}(t)}\quad {t}}}} = {1500000{\int_{- {.01}}^{.01}{{{t\varphi}(t)}\quad {t}}}}}$

Unfortunately, directly detecting the phase as a function of time inorder to apply such a weighting is just as impractical for averagingphase rate as for averaging phase, because of the noise that resultswhen phase is measured in a wide bandwidth to produce φ(t).

Similarly applying the ramp weighting for Doppler-shift frequencymeasurements to the signal-vector samples, on the other hand, yields thevector velocity of the signal, not the scalar frequency-shiftmeasurement desired.

In the present invention, low-noise measurements of phase rate orfrequency shift is preferably based on the commutative property oflinear processing. Linear operations to the signals can be applied inany order. The nonlinearity of phase detection, which must occur at somepoint in the processing, is tolerated by invoking the approximation thatit behaves like a linear process when the variations in phase are small.This condition is satisfied when the phase change and the noiseamplitude during the measurement are both small compared to one radian.The detection threshold may then be thought of as the signal-to-noiseratio for which this approximation is no longer valid.

The basic linear processing of the present invention comprisescorrelation of the ramp weighting function with the signal samples. Thecomplementary linear operations of integration and differentiation areinserted into such processing without changing the overall result. Theintegration operation is applied to a ramp-like frequency-measurementweighting function to create a parabolic “hump” weighting:$s_{h} = {{\int_{{- T}/2}^{T/2}{\left\lbrack {\int_{{- T}/2}^{t}{\frac{12t}{T^{3}}{t}}} \right\rbrack {s(t)}\quad {t}}} = {{\frac{1.5}{T}{\int_{{- T}/2}^{T/2}{\left\lbrack {\frac{4t}{T^{2}} - 1} \right\rbrack {s(t)}\quad {t}}}} = {75{\int_{- {.01}}^{.01}{\left( {{10000t^{2}} - 1} \right){s(t)}\quad {t}}}}}}$

The differentiation is applied to the resulting average. The net effectis no change to the result, under the assumption of linearity.

After averaging of the signal vector with the hump weighting, the noisewill be substantially reduced. Hump weighting can be seen to be onlyabout 0.8 decibels noisier than uniform weighting by averaging thesquare of the ratio of the weighting functions over the measurementtime:${\frac{1}{T}{\int_{{- T}/2}^{T/2}{\left( \frac{1.5}{T} \right)^{2}\left( \frac{4t^{2}}{T^{2}} \right)^{2}T^{2}\quad {t}}}} = 1.2$

This point in the processing is an opportune time to do the detectionsince the high signal-to-noise ratio validates the assumption oflinearity. The resulting phase measurements from two successivehump-weighted averages are differenced to obtain the phase rate. Suchhybrid vector-scalar processing approaches the optimum processing oflow-noise short-term phase measurements, and continues to do well evenin high-noise, high-rate situations.

The absolute limit for this kind of velocity measurement occurs when thehump integral goes to zero. The amplitude of the hump integral, as afunction of Doppler shift ω, is,s_(h) = 75∫_(−.01)^(.01)(10000t² − 1)cos (ωt)  t = −3000000ω⁻³sin (.01ω) + 30000ω⁻²cos (.01ω)

The frequency of the first null in this function is found by setting itto zero and solving for ω:

tan (0.01ω)=0.01ω

ω=449.3=2φ71.5 (Hz).

The three-decibel point is at 28.89 hertz.

The detection threshold is defined as equal signal and noise power atthe detection bandwidth. The following scaling of the integral of thesquare of the hump weighting function results in a phase noise of about{square root over (0.5)} radian, which corresponds to equal signal andnoise power: σ_(φ)² = 46.875∫_(−.01)^(.01)(1 − 10000t²)²  t = .5

The same scaling, but with the integrand modified by the square of theratio of the weighting functions for the ramp and the hump, gives thevelocity noise:$\sigma_{\omega}^{2} = {46.875{\int_{- {.01}}^{.01}{\frac{1500000^{2}}{75^{2}}t^{2}\quad {t}}}}$σ_(ω) = 111.8 = 2π ⋅ 17.8Hz

This is less than one quarter the 71.5 hertz range of the velocitymeasurement technique. If the signal remains at the detection threshold,the velocity noise can be expected to exceed four standard deviationsand cause a loss of frequency lock about once every half hour, at tenmeasurements per second.

To implement the Doppler-frequency measurement, navigation computer 30computes hump-weighted sums of n−1 consecutive signal samples. Itmeasures the phase angles of two such successive vector sums spanningthe n samples of one twenty-millisecond data-bit time. The phase rate orfrequency shift is then the difference between the angles, divided bythe sampling interval:$\omega = \frac{{{angle}\left( {hump}_{i} \right)} - {{angle}\left( {hump}_{i - 1} \right)}}{\Delta \quad t}$

Navigation computer 30 may compute these hump-weighted sums explicitlyor, to reduce the computational workload, by adding to the previous humpsum the ramp-weighted sum of the n most-recent signal samples, whichconverts the previous hump to the present hump:

hump_(i)=hump_(i−1)+ramp_(i)

To prevent accumulation of computational errors, these and thecalculations that follow must be done in exact integer arithmetic. Thisramp-weighted sum may be computed explicitly at each iteration or invarious ways that further reduce the computational workload. One methodis, like the hump computation, to compute each ramp by adding to theprevious ramp a quantity to convert it to the present ramp:

ramp_(i)=ramp_(i−1) −k sum_(i)+(k/2)(n+1)(signal_(i)+signal_(i−n))

where k must be even for even n but may have any integer value for oddn. The overall scaling of the weighting process is immaterial, since thephase-angle change does not depend on it. The variable sum is maintainedin a similar manner as the uniformly weighted sum of the n+1 most-recentsamples:

sum_(i)=sum_(i−1)+signal_(i)−signal_(i−n−1)

Another efficient way of computing the ramp-weighted sum is as acombination of an asymmetrical ramp and a uniform sum:

ramp_(i) =k _(i) aramp−(k/2)(n−1)sum_(i)

 aramp_(i)=aramp_(i−1)−sum_(i) +n signal_(i)

sum_(i)=sum_(i−1)+signal_(i)−signal_(i−n)

Subscripts are shown on all vector variables for clarity to relate themto the iteration scheme; but, since most are simply incremented at eachiteration, an array of vectors is needed only for the signal samples.

Although the present invention uses a frequency measurement to track thesignal, it is not restricted from also making phase measurements. Up toalmost two G, it can measure the phase shift unambiguously under theassumption of no acceleration since the last measurement, as alreadydescribed. However, by using the present frequency measurement incombination with the previous one or ones it can estimate accelerationsince the last measurement and therefore track phase without ambiguityup to much higher accelerations.

The frequency measurement method of the present invention can also beused to advantage to improve the dynamic tracking of non-sequencingreceivers by providing a direct short-term measurement of Doppler shift.

Although the present invention has been described in terms of thepresently preferred embodiment, it is to be understood that thedisclosure is not to be interpreted as limiting. Various alterations andmodifications will no doubt become apparent to those skilled in the artafter having read the above disclosure. Accordingly, it is intended thatthe appended claims be interpreted as covering all alterations andmodifications as fall within the true spirit and scope of the invention.

What is claimed is:
 1. A navigation satellite receiver comprising: afirst receiver channel or set of channels connected to a radio antennafor demodulating a navigation-data modulation on any of a plurality ofradio transmissions from a corresponding plurality of navigationtransmitters and for acquiring signals from additional transmitters; asecond receiver channel or set of channels connected to a radio antennafor sequentially tracking a plurality of radio transmissions from acorresponding plurality of transmitters; means for sequencing the secondreceiver channel or channels for collecting a period of measurementssequentially from each of said plurality of transmitters that coincideswith the bit times of said navigation data in which said transmissionsare coherent; and a computing means for measuring the Doppler shift orfrequency error.
 2. The receiver of claim 1, further comprising:computing means for tracking the phase of the signal vectormeasurements, wherein the phase ambiguity caused by large accelerationsis resolved by using the successive Doppler-shift measurements toestimate and remove the effect of the acceleration.
 3. The receiver ofclaim 1, wherein: the sequencing means switches after each data-bit timeto the transmission whose next data transition occurs soonest, so thatthe aggregate time wasted waiting for each complete data-bit time tobegin on the next transmission can never exceed the duration of one databit over the complete sequencing cycle.